Recently, a novel architecture of receiver is being used increasing popular in the radio communication field. The basic design concept of such receiver architecture is to make analog to digital converters (ADC) close to radio frequency (RF) receiving antennas as much as possible so as to perform analog to digital conversion directly on the radio frequency signal, and then programmable digital signal processing units process received signals. Owing to the characteristics of flexibility, low cost and easy to integrate of digital signal processing, such method is applicable for many kinds of communication protocol, and thus facilitates technical upgrading.
Nevertheless, the above-mentioned type of receiver will be taken herein as an example to illustrate the problems such receiver is encountering.
Firstly, a user signal (with the bandwidth of B) is represented by two orthogonal components, that is, I(t)+jQ(t), then the radio frequency signal with the carrier frequency of fc which is orthogonally modulated by the user signal could be represented as:S(t)=I(t)cos(ωct+φ)−Q(t)sin(ωct+φ)  (1)
Wherein ωc=2πfc, representing the circular frequency of the carrier, and φ is the initial phase of the carrier.
S(t) could be further represented by two band-pass components S′(t) and S″(t) whose central frequencies are fc and −fc, respectively:
                                          S            ′                    ⁡                      (            t            )                          =                              1            2                    ⁢                      {                                          [                                                                            I                      ⁡                                              (                        t                        )                                                              ⁢                                          cos                      ⁡                                              (                        φ                        )                                                                              -                                                            Q                      ⁡                                              (                        t                        )                                                              ⁢                                          sin                      ⁡                                              (                        φ                        )                                                                                            ]                            +                              j                ⁡                                  [                                                                                    I                        ⁡                                                  (                          t                          )                                                                    ⁢                                              sin                        ⁡                                                  (                          φ                          )                                                                                      +                                                                  Q                        ⁡                                                  (                          t                          )                                                                    ⁢                                              cos                        ⁡                                                  (                          φ                          )                                                                                                      ]                                                      }                    ⁢                      ⅇ                          j              ⁢                                                          ⁢                              w                c                            ⁢              t                                                          (        2        )                                                      S            ″                    ⁡                      (            t            )                          =                              1            2                    ⁢                      {                                          [                                                                            I                      ⁡                                              (                        t                        )                                                              ⁢                                          cos                      ⁡                                              (                        φ                        )                                                                              -                                                            Q                      ⁡                                              (                        t                        )                                                              ⁢                                          sin                      ⁡                                              (                        φ                        )                                                                                            ]                            -                              j                ⁡                                  [                                                                                    I                        ⁡                                                  (                          t                          )                                                                    ⁢                                              sin                        ⁡                                                  (                          φ                          )                                                                                      +                                                                  Q                        ⁡                                                  (                          t                          )                                                                    ⁢                                              cos                        ⁡                                                  (                          φ                          )                                                                                                      ]                                                      }                    ⁢                      ⅇ                                          -                j                            ⁢                                                          ⁢                              w                c                            ⁢              t                                                          (        3        )            
The frequency spectrum characteristics thereof are as shown in FIG. 1, and it can be seen that the signal bandwidths of S′(t) and S″(t) are completely the same.
To avoid frequency spectrum aliasing when performing band-pass sampling on the radio frequency signal, a clock signal with the frequency of
      f    s    =                    f        c            N        >    B  could be selected, and the sampled signal frequency spectrum is equivalent to that the original RF signal frequency spectrum (as shown in FIG. 1) periodically extended in the frequency spectrum domain, taking the sampling frequency fs as the period, as shown in FIG. 2.
It can be seen from FIG. 2 that when the frequency spectrum is periodically extended, the high order frequency spectrum components of S′(t) and S″(t) are superposed with each other at frequencies which are integral multiple of the sampling frequency. Therefore, there is a superposed frequency component with the bandwidth of B at the zero frequency, for baseband signal processing subsequently. The time domain representation of the signal having the zero frequency as its center (i.e., the carrier frequency thereof is zero) could be calculated from equations (2) and (3), that is, I(t)cos(φ)−Q(t)sin(φ). Due to the frequency spectrum aliasing, the zero carrier signal is in fact a linear combination of the orthogonal user signals I(t) and Q(t).
In order to separate the orthogonal user signals I(t) and Q(t), two clock signals having the same frequency but different phases are used to perform two-path band-pass sampling on the RF signal, thereby to obtain the linear combination of two orthogonal user signals that are different from each other, then the I(t) and Q(t) of the user signal could be obtained through separation process.
According to the above principle, a solution is put forward in the Chinese application for patent for invention titled “Band-pass sampling receiver and the sampling method thereof” with the number of 200310122502.3 filed by the applicant of Koninklijke Philips Electronics N. V. on Dec. 5, 2003, and the contents disclosed in this application will be introduced herein by insertion. The architecture of the band-pass sampling receiver provided by this patent application is as shown in FIG. 3, wherein the sampling clock frequencies of two ADC 710 and 711 are both 1/N of the carrier frequency of the RF signal, but there is a fixed relative time delay τ between two sampling clocks CLK1 and CLK2, while
      τ    ⁢          <<              1        B              ,thus making the sampling points of the two paths of clock signals to have different carrier phases. Therefore, two different digital sequences are obtained after analog to digital conversion. Necessary digital signal processing is performed so as to separate the two orthogonal components and recover the desired user signal subsequently.
The above solution is an effective sub-sampling solution under ideal conditions owing to its relatively low sampling frequency and putting main signal processing on the more flexible digital domain.
However, under some circumstances, there will be stronger DC drift and intermodulation component caused by non-ideal circuits, and such interference is usually hard to accept and need to be removed by filters or by compensation algorithm. For communication systems like IS-95, CDMA2000 and UMTS systems, energy of the useful signal thereof is distributed in a broad frequency domain, while the interference of DC drift and intermodulation component, etc., are within a narrow frequency domain near the zero frequency, in which energy of the signal takes only a very small share of the total signal energy, thus the interference could be filtered out through the digital high-pass filter without imposing great influence on the performance of the useful signal. However, for the communication systems such as GSM and Bluetooth, since energy of the useful signal thereof is mainly concentrated in a narrow range near the zero frequency domain, a lot of useful signals will be lost when filtering the interference near the zero frequency domain through the digital high-pass filter; so such method is not suitable for the zero intermediate frequency band-pass sampling receiver.
Therefore, it becomes a pending problem as to how to improve the structure of the existing receiver and the sampling method thereof so as to effectively filter out the interference of DC drift and intermodulation component, etc., and thus to make it applicable to more kinds of communication systems.